A Compact, Ultra-Wideband, Transformer-Based Quadrature Signal Generation Network in 45 nm CMOS SOI for 5G Applications

Abstract

In this article, we present an ultra-wideband, fully-differential quadrature signal generation network for 5G applications. The ultra-wideband network is composed of a passive balun and cascaded transformer-based quadrature hybrid. Passive balun converts a single-ended signal to differential with minimum insertion loss, and transformer-based quadrature hybrids are cascaded to suppress I/Q imbalance over an ultra-wide bandwidth. The coupling coefficient of the transformer-based quadrature hybrid is enhanced by adopting vertically stacked multiturn transformer topology to extend operation bandwidth and reduce passive loss and chip area. A novel layout and signal routings are proposed to reduce passive loss, undesired magnetic coupling and I/Q imbalance, making meander lines for phase matching unnecessary. The proposed ultra-wideband quadrature signal generation network is designed in GlobalFoundries 45 nm CMOS SOI process with a core area of 845 μm $\times$ 495 μm. The output I/Q magnitude mismatch is less than 0.5 dB from 16 to 60 GHz, and phase mismatch is less than 2° from 16.5 to 54.7 GHz. The input return loss is lower than −10 dB from 22 to 45 GHz, and signal loss varies from 5.74 to 7.4 dB (including 1:2 power splitting and loss from passive balun). The effective image rejection ratio (IRR) is calculated based on I/Q mismatch and is higher than 40 dB from 21.5 to 53.5 GHz.

1. Introduction

The development of modern mobile communication has entered the 5G millimeter-wave era. The new standard of 5G, which operates at 24–29 GHz and 37–43 GHz, provides higher bandwidth and Gb/s data rate, enabling various applications, such as wireless system and mobile phone user communication. Growing demands for 5G mobile equipment increase the research interest in advanced millimeter-wave 5G circuit design and block integration [1,2,3,4,5,6,7,8,9,10,11,12,13,14,15,16].

Generally, a 5G base station is arranged to cover a 100–300 m radius [11], which means that millimeter-wave data links could suffer from a large space loss and atmospheric attenuation such a long distance. In order to improve millimeter-wave data-link performance and increase spatial diversity, phased-array, as well as multiple-input and multiple-output (MIMO) architectures, are widely proposed and extensively utilized. Several narrow-band low noise amplifiers (LNA) and phased-array systems have been demonstrated for 5G application, but they actually operate either at 28 GHz or 39 GHz [3,4,5,6,7,8,9,10]. They fail to cover the whole 5G standard frequency band with a single system. Multiple phased-arrays are therefore cointegrated to cover the whole 5G frequency band, which would substantially increase chip size and power consumption. Therefore, it is desirable to design a phased-array system covering the whole 5G frequency band (23–44 GHz). Recently reported wideband phased-array beamformers, LNAs, power amplifiers (PA) and T/R switches have achieved decent performance and great success [12,13,14,15,16], making it possible to support 23–44 GHz operation using a single system.

In 5G wideband receivers, quadrature signals have been widely used to perform beamforming (together with vector modulator) and avoid image jamming. Figure 1 demonstrates one such application in image rejection. A single-ended LO signal is first converted to a differential signal by using a passive balun and fed to an I/Q signal generation network. Then, a differential quadrature LO signal down-converts the RF signal in mixers. After the down-conversion operation, the differential quadrature signal is amplified in an IF variable-gain amplifier (VGA) and then combined.

IRR highly depends on the quality of the quadrature signal, i.e., phase mismatch and magnitude mismatch. However, it is a great challenge to generate a perfectly matched quadrature signal that supports wideband 5G applications. RC-CR polyphase filters (PPFs) [17] are popular for low-RF frequency bands due to their relatively compact size and simplicity. The quadrature signal generation bandwidth could be extended by cascading multiple stages of RC-CR PPFs. However, in the 5G frequency band, the resistive components in the signal path attenuate RF signal substantially. Moreover, RC-CR PPFs are sensitive to process mismatch [16] and vulnerable to load impedance variation. A quadrature all-pass filter (QAF) network is another possible solution to generate an I/Q signal. In QAF networks, I/Q errors due to load capacitance could be suppressed by adding series-resistive components at the cost of higher signal loss. A inductive compensation network can also be added at the output to reduce its load sensitivity and achieve voltage peaking [16] but with a substantial increase in chip area. Moreover, the I/Q signal quality of QAF networks (roughly 5° phase imbalance and 0.5 dB magnitude imbalance) cannot satisfy the requirement of an image rejection ratio (IRR) higher than 30 dB (IRR > 30 dB typically requires the phase and magnitude mismatch to be <3° and <0.25 dB, respectively [12]). Without I/Q signal generation, a front-end with IRR greater than 75 dB was achieved in [15] by properly designing the frequency plan and adopting a high-pass filter in LNA but at the expense of higher noise figure (NF) and design complexity.

In [18], ultra-wideband quadrature signal generation was achieved with state-of-art I/Q balance performance by cascading transformer-based quadrature hybrids (without resistive components). Compared with a single-stage transformer-based quadrature hybrid [19,20], the I/Q imbalance of the quadrature signal generation network in [18] was highly suppressed, and bandwidth was extended significantly, making ultra-wideband, precise and low-loss I/Q signal generation possible. However, relatively long meander lines (for phase matching) and signal routes introduce extra loss and increase chip area and design complexity. Moreover, coupling coefficient k could be further enhanced to reduce loss and minimize layout footprint.

In this paper, a compact and ultra-wideband quadrature signal generation network (shown in Figure 1) is proposed for 5G applications. The wideband passive balun is designed to achieve minimum loss, and the I/Q generation network is optimized for wideband operation, low loss and compact area.

This article is organized as follows. In Section 2, we introduce the characteristics of the GlobalFoundries 45 nm CMOS SOI process. In Section 3, we present the proposed transformer-based quadrature hybrid, its detailed operation principles and circuit implementation. In Section 4, we describe the design details of the proposed ultra-wideband I/Q signal generation network. In Section 5, we present the simulation results and a performance comparison with reported I/Q generation schemes. Finally, in Section 6, we draw conclusions.

2. Technology

The ultra-wideband quadrature signal generation network described in this paper is designed in the GlobalFoundries 45 nm CMOS SOI process. The process provides seven copper metal layers and one aluminum top metal layer, which is generally used to implement measurement PADs, wire bonding and high-density MIM capacitors. A cross-section view of the metal stacks is shown in Figure 2, together with the thickness of each metal layer. As an important characteristic of CMOS SOI technology, one buried oxide layer is located beneath the metal stacks, which could reduce the loss from silicon substrate compared with standard CMOS technology. Owing to the top-three thick metal layers (OA, OB and LD), a 300 pH spiral inductor designed in OB layer could achieve a simulated Q of 20 at 30 GHz. Figure 2 also shows a coplanar waveguide (CPW) transmission line, where signal is propagated in the OB layer and M1-LD could be stacked as ground wall to reduce magnetic coupling between neighboring passive components.

In addition, the OA and OB layers have the same thickness and sheet resistance, with the potential to provide similar magnitude and phase performance in a hybrid quadrature coupler. In this way, the quadrature signal generation network proposed in this article is mainly built in OA and OB layers.

3. Transformer-Based Quadrature Hybrid

A quadrature hybrid is a four-port (input, through, coupled and isolate) network composed of two transmission lines, as shown in Figure 3a. When the input port is connected to a signal source, mutual magnetic coupling occurs between two transmission lines, known as inductive coupling. Capacitive coupling is provided by the capacitor between two transmission lines. Inductive and capacitive coupling are both utilized in the quadrature hybrid to achieve matched quadrature signals at the through port and coupled port.

Applying even-mode and odd-mode analysis to the quadrature hybrid, as shown in Figure 3b,c, the voltage and current equation of transmission line can be expressed as [21]:

$$V(z)=V^{+}\cdot (e^{-j\beta l}+\Gamma e^{j\beta l})$$

(1)

$$I(z)={\frac{V}{Z_c}}^{+}\cdot (e^{-j\beta l}-\Gamma e^{j\beta l})$$

(2)

where $Z_c$ is the characteristic impedance of the transmission line; $\beta$ is the phase shift of a unit-length transmission line; $l$ is the electrical length of the transmission line; and $\Gamma$ is the reflection factor, which is expressed as:

$$\Gamma =\frac{Z_0-Z_c}{Z_0+Z_c}$$

(3)

When the even-mode and odd-mode signals are applied to the transmission lines, the following equations are satisfied due to symmetry:

$$V_1^e=V_3^e$$

(4)

$$V_2^e=V_4^e$$

(5)

$$V_1^o=-V_3^o$$

(6)

$$V_2^o=-V_4^o$$

(7)

where $V_1^e$, $V_2^e$, $V_3^e$ and $V_4^e$ are the node voltages in even-mode at nodes 1, 2, 3 and 4, respectively; $V_1^o$, $V_2^o$, $V_3^o$ and $V_4^o$ are the node voltages in odd-mode at nodes 1, 2, 3 and 4, respectively. Therefore, the input impedance in even-mode and odd-mode can be calculated as:

$$Z_{in}^e=Z_{0e}\frac{Z_0+j{Z}_{0e}\tan \beta l}{Z_{0e}+j{Z}_0\tan \beta l}$$

(8)

$$Z_{in}^o=Z_{0o}\frac{Z_0+j{Z}_{0o}\tan \beta l}{Z_{0o}+j{Z}_0\tan \beta l}$$

(9)

where $Z_0=\sqrt{Z_{0e}\cdot Z_{0o}}$. With the voltage relationship, we have:

$$V_1^e=V_s\frac{Z_{in}^e}{Z_{in}^e+Z_0}$$

(10)

$$I_1^e=\frac{V_s}{Z_{in}^e+Z_0}$$

(11)

$$V_1^o=V_s\frac{Z_{in}^o}{Z_{in}^o+Z_0}$$

(12)

$$I_1^o=\frac{V_s}{Z_{in}^o+Z_0}$$

(13)

where $I_1^e$ and $I_1^o$ are the current flows into node 1 in even-mode and odd-mode, respectively. By substituting (8)–(9) into (10)–(13) and taking the superstition method into consideration, the voltage at each port ($V_1$, $V_2$, $V_3$ and $V_4$) can be expressed as:

$$V_1=V_1^e+V_1^0=V_s$$

(14)

$$V_2=V_2^e+V_2^o=V_s\frac{\sqrt{1-C^2}}{\sqrt{1-C^2}\cos \beta l+j\sin \beta l}$$

(15)

$$V_3=V_3^e+V_3^o=V_s\frac{jC\tan \beta l}{\sqrt{1-C^2+j\tan \beta l}}$$

(16)

$$V_4=V_4^e+V_4^o=0$$

(17)

where $C=(Z_{0e}-Z_{0o})/(Z_{0e}+Z_{0o})$ is the voltage coupling coefficient. When the length of the transmission line is equal to $\lambda /4$, namely $\beta l=\pi /2$, (14)–(17) are simplified as:

$$\frac{V_2}{V_s}=-j\sqrt{1-C^2}$$

(18)

$$\frac{V_3}{V_s}=C$$

(19)

When $C=\sqrt{2}/2$, perfect quadrature signals with same magnitude and 90° phase difference are generated at the coupled port and through port.

The transmission line quadrature hybrid can achieve a matched I/Q signal at the output ports, but the chip area occupied by the $\lambda /4$ transmission line is considerably large, especially in the low-RF frequency band. However, a transmission line hybrid could be realized by transformers with a compact chip area and similar I/Q balance performance. Figure 4 shows the equivalent lumped-element circuit of a fully differential transformer-based quadrature hybrid. Leakage and series losses of inductors are represented by an inductor in parallel with the resistor. Similar to the transmission line quadrature hybrid, a transformer-based quadrature hybrid has four ports: input, coupled (CPL), through (THRU) and isolate (ISO). It also utilizes inductive coupling (provided by transformers 1 and 2) and capacitive coupling (provided by the parasitic capacitor Cm) to generate I/Q signal at the through port and coupled port. The isolate port is connected to Z0 ($100\Omega$), which is the matched impedance of the port.

Figure 5 shows the 3D EM model of the proposed quadrature hybrid, wherein two transformers are folded and stacked so that the quadrature hybrid occupies only one inductor footprint. The inductors on the routes from input port to through port are designed in the OB layer (purple spiral polylines), inductors on the routes from coupled port to isolate port are designed in the OA layer (orange spiral polylines) and crossovers are designed in the LD and UA layers. Ground walls composed of M1-LD layers are placed around the quadrature hybrid to serve as a signal return path and to improve magnetic coupling isolation. With a proper design, it will produce a 90° phase difference between coupled port and through port at a given frequency. A 100 Ω resistor is placed at the isolate port to avoid signal reflection and improve impedance matching. The whole layout is symmetric in order to reduce magnitude and phase mismatch, except crossovers. The width of all metal polylines is 6μm, and the space between them is 10 μm. The layout size of the proposed quadrature hybrid is only 220 μm $\times$ 160 μm.

The quadrature hybrid proposed in this article has the following advantages over the hybrids presented in [18,19]: (1) a vertically stacked, multiturn transformer topology is adopted, enhancing the magnetic coupling coefficient (0.89) between transformers 1 and 2 so that a more compact layout, lower passive loss and a wider quadrature balance bandwidth can be achieved; (2) signal routes from input port to through port and from coupled port to isolate port are properly designed in OA and OB layers with same thickness and sheet resistance, which could reduce intrinsic magnitude and phase imbalance; (3) the coupling capacitor Cm shown in Figure 5 is absorbed in parasitic capacitance between metal spiral polylines. The absence of coupling capacitor results in a smaller chip area and removes modeling uncertainty. The advantages are justified by the following simulation results.

Figure 6 presents the simulation results of the proposed quadrature hybrid, the port definition of which is shown in Figure 5. Figure 6a shows that the return loss of input, through and coupled ports are all below −10 dB from 10 GHz to 60 GHz; the insertion loss of coupled and through ports are better than −5 dB (including 3 dB from 1:2 power dividing) from 20 GHz to 60 GHz; and the insertion loss is 3.6 dB at 29 GHz (also including 3 dB from power splitting), which is the perfect magnitude matched point. Figure 6b shows the magnitude and phase imbalance of the quadrature hybrid. The magnitude mismatch is below 2 dB from 20.5 GHz to 60 GHz, and the phase mismatch is below 5° from 10 GHz to 45 GHz, which demonstrates the wideband operation capability of the proposed quadrature hybrid.

4. Transformer-Based Ultra-Wideband I/Q Signal Generation

As demonstrated in the previous section, our proposed quadrature hybrid has the capability to generate a fully differential I/Q signal over a wide bandwidth. However, due to limited perfect phase balance bandwidth, it fails to satisfy a wideband frontend targeted at the 5G frequency band. For example, the magnitude mismatch is still below 1 dB from 37 GHz to 43 GHz, whereas the phase mismatch is larger than 3°, which means that the front end cannot achieve an IRR > 30 dB (phase mismatch <3° with magnitude mismatch <0.25 dB) with only one quadrature hybrid.

However, the I/Q imbalance could be considerably suppressed by cascading quadrature hybrid unit stages after a quadrature hybrid [18]. The block diagram is depicted in Figure 7. The quadrature hybrid unit stage is composed of two identical quadrature hybrids in a symmetric manner and an eight-to-four signal combination route, so the unit stage has two differential input ports and four differential output ports. The operation principles of transformer-based quadrature hybrid unit stage are demonstrated as follows. The transformer-based quadrature hybrid first converts a fully-differential signal into two fully-differential quadrature signals at the output ports. Two pairs of differential signals are then fed to the first transformer-based quadrature hybrid unit stage. As a result, two differential signals with 90° phase difference are generated at the output of the first unit stage after signal combination and connected to the second unit stage. The common-mode and differential-mode characteristics of the transformer-based quadrature hybrid unit stage will suppress I/Q imbalance that originates from the previous circuits [18]. Finally, two differential signals with high-quality quadrature performance occur at the output of the nth transformer-based quadrature hybrid unit stage.

Figure 8 shows the signal combination mechanism in the transformer-based quadrature hybrid unit stage. Two quadrature differential signals are fed to the unit stage after the quadrature hybrid. A pair of differential signals is connected to quadrature hybrid A (represented by black lines in the upper part of the quadrature hybrid unit stage in Figure 7) in the unit stage, whereas another signal is fed to quadrature hybrid B (represented by gray lines in bottom part of the quadrature hybrid unit stage in Figure 7). Therefore, at the output of the unit stage, there are total eight output signals but with only four different phases, which is also shown in Figure 8. In this way, eight output signals can be combined and converted into two quadrature differential signals. For example, the output 0° signal is composed of an input 0° signal passing through the THRU port in quadrature hybrid A and an input −90° signal passing through the CPL port in quadrature hybrid B. The signal combination operation of the transformer-based quadrature hybrid unit stage can average out existing phase/magnitude imbalance from previous stages.

The 3D EM model of proposed ultra-wideband two-stage cascaded transformer-based quadrature hybrid (composed of a transformer-based quadrature hybrid and one transformer-based quadrature hybrid unit stage) is depicted in Figure 9, together with circuit blocks and port notations. Ground walls from M1 to LD are placed around each block to improve the magnetic coupling isolation. The input port is driven by a 100 Ω differential port, and two output ports are each terminated with a 100 Ω differential port.

In the transformer-based quadrature hybrid unit stage presented in [12,18], the differential signal routes between the quadrature hybrid and unit stage are quite long, which increases the chip area and design complexity. Additionally, the meander lines for phase matching adopted in [18] increase chip area and introduce extra passive loss. In this article, ports, unit stages, signal routing lines and signal combination routes are carefully arranged. Therefore, the length of signal routing lines is minimized to reduce I/Q imbalance, and a meander transmission line is not necessary. Moreover, in order to reduce phase/magnitude mismatch generated by undesired magnetic coupling, ground walls from M1 to LD are placed around each quadrature hybrid. In this way, the quadrature hybrids can be arranged closer to each other to further reduce chip area and length of signal routing lines.

Before the cascaded transformer-based quadrature hybrid, a wideband passive balun converts the single-ended signal into a differential signal, as shown in Figure 10. The primary and secondary coils of the balun are implemented using OB and OA layers, respectively. The coupling factor between the primary and secondary coils is 0.72, and the core balun size is 186 μm $\times$ 176 μm. Series and shunt capacitors are added at the input and output terminals of balun to achieve wideband impedance matching. The simulated balun insertion loss is <2 dB, with an input return loss <−10 dB at 22.5–49 GHz, as shown in Figure 11.

5. Simulation Results

The proposed ultra-wideband I/Q signal generation network is designed in GlobalFoundries 45 nm CMOS SOI process. Full-wave electromagnetic simulator Ansys and Cadence Virtuoso were used to simulate the circuit. The layout of I/Q signal generation network is shown in Figure 12, with a core chip area of 845 μm $\times$ 495 μm (including input passive balun), demonstrating a compact layout footprint.

Figure 13 shows the typical S-parameter results of the proposed I/Q signal generation network. The input return loss is lower than −10 dB from 22 GHz to 45 GHz, which validates wideband operation capability of the proposed network. The minimum passive insertion loss of the I/Q generation network is only 5.74 dB at 25 GHz and increases to 7.4 dB at 45 GHz (including 3 dB contribution from 1:2 power splitting and loss from passive balun). Therefore, the passive loss of the core cascaded transformer-based quadrature hybrid is only 1.9 dB at 25 GHz and increases to 2.9 dB at 45 GHz. S-parameter results validate the design methodology of our proposed I/Q generation network for wideband and low-loss characteristics.

For an I/Q signal, the phase imbalance is defined as:

$$\mathrm{Phase} \mathrm{Imbalance}=\left|{ 90}^{\mathrm{o}}-\left|\mathrm{Phase}( \mathrm{I}\_\mathrm{Output} , \mathrm{Input} )-\mathrm{Phase}( \mathrm{Q}\_\mathrm{Output} , \mathrm{Input} )\right|\right|$$

The magnitude imbalance is defined as:

$$\mathrm{Magnitude} \mathrm{Imbalance}=\left|\mathrm{dB}( \mathrm{I}\_\mathrm{Output} , \mathrm{Input} )-\mathrm{dB}( \mathrm{Q}\_\mathrm{Output} , \mathrm{Input} )\right|$$

The phase/magnitude imbalance of the proposed I/Q generation network is depicted in Figure 14. The phase imbalance is lower than 2° from 16.5 GHz to 54.7 GHz and lower than 5° from 12 GHz to 59 GHz. The magnitude imbalance is lower than 0.5 dB at 16–60 GHz and lower than 1 dB from 13 GHz to 60 GHz. The phase/magnitude imbalance performance of our proposed I/Q generation network demonstrates its ability to output ultra-precise, fully-differential quadrature signals over an extremely wide bandwidth.

The quality of I/Q signal (including phase and magnitude imbalance) is often evaluated using IRR [12], which can be calculated as:

$$\mathrm{IRR}=\frac{{(1+\epsilon )}^2+2(1+\epsilon )\cos (\Delta \theta )+1}{{(1+\epsilon )}^2-2(1+\epsilon )\cos (\Delta \theta )+1}$$

where $\epsilon$ is the magnitude imbalance, and $\Delta \theta$ is the phase imbalance. When $\epsilon$ and $\Delta \theta$ are both small, IRR can be simplified as [22]:

$$\mathrm{IRR}\approx \frac{4}{{\epsilon }^2+\Delta {\theta }^2}$$

(20)

The IRR of the proposed I/Q signal generation network is calculated and plotted in Figure 15, together with the IRR of transformer-based quadrature hybrid. Both circuits are loaded with 100 Ω resistor at differential I/Q output ports. The transformer-based quadrature hybrid can only achieve >30 dB IRR from 28 GHz to 30 GHz and >20 dB IRR from 22 GHz to 50 GHz, which indicates that the quadrature hybrid cannot generate a precise I/Q signal over a wide bandwidth, whereas our proposed I/Q generation network achieves >30 dB IRR from 16.5 GHz to 56 GHz and >40 dB IRR from 21.5 GHz to 53.5 GHz. Compared with one-stage quadrature hybrid, the proposed I/Q generation network can support an ultra-high IRR over a wide bandwidth without calibration.

Table 1. Performance summary and comparison.

Reference	Type	Frequency Range (GHz)	Gain (dB)	I/Q Mag/Phase Error (dB/deg)	IRR (dB)	Chip Size (μm2)
This Work a	Passive balun + two-stage cascaded transformer-based quadrature hybrid	21.5–53.5	−2.74 (−1.9 c)	$0.13/1.2{}^{\circ }$	>40	845 × 495 d
[12] b	Two-stage transformer-based differential I/Q network	25–50	−2.1	$0.15/1.8{}^{\circ }$	>35.2	330 × 425
[16] a	Quadrature all-pass filter	17–41.3	+3	$1/5{}^{\circ }$	-	240 × 410
[17] b	Three-stage RC-CR polyphase filter	2.5–10	−10	-	>35	-
[18] b	Three-stage transformer-based polyphase network	3.7–22.5	−3.65	$0.5/2{}^{\circ }$	>30	772 × 925
[19] b	Folded transformer-based quadrature hybrid	4.75–5.41	−0.82	$0.5/3.8{}^{\circ }$	-	260 × 260
[20] b	Single-ended transformer-based quadrature hybrid	1.95–2.05	−1.7	-	>30	390 × 350

^a simulation, ^b measurement, ^c without passive balun, ^d including passive balun.

summarizes the performance of our proposed ultra-wideband quadrature signal generation network and presents a comparison with state-of-the-art quadrature generation schemes. Figure 16 presents a comparison in terms of figure. The quadrature all-pass filter in [16] and the one-stage transformer-based quadrature hybrid in [19,20] can only provide a matched I/Q signal over a narrow bandwidth, limiting their applications in wideband 5G transceivers. An RC-CR filter can generate an I/Q signal with a high IRR over wide bandwidth but with substantially high signal loss and vulnerability to process variation. The network proposed in [18] can provide I/Q signal with an IRR > 30 dB over a wider bandwidth. However, the scheme, which is composed of three stages of transformer-based poly phase network, results in extra signal loss and chip area. Compared with wideband I/Q signal network presented in [12], the ultra-wideband quadrature signal generation network proposed in this paper can generate matched I/Q signal with a higher IRR over a wider bandwidth.

6. Conclusions

In this paper, an ultra-wideband, fully-differential quadrature signal generation network is described and analyzed in detail. In order to generate an I/Q signal with perfectly matched I/Q balance in the entire 5G frequency band, a transformer-based quadrature hybrid is cascaded to suppress I/Q imbalance. Multiturn transformers are stacked vertically to enhance the coupling coefficient of the transformer-based quadrature hybrid. The layout of the cascaded transformer-based quadrature hybrid is carefully arranged to minimize intrinsic passive loss, as well as undesired I/Q imbalance. Magnetic coupling is minimized by adding ground wall shield between neighboring circuit blocks; thus, the quadrature hybrids can be placed in a compact manner to save chip area. Designed in GlobalFoundries 45 nm CMOS SOI process, our proposed ultra-wideband quadrature signal generation network shows excellent I/Q balance performance. The output I/Q magnitude mismatch is lower than 0.5 dB at 16 GHz –60 GHz, and the phase mismatch is lower than 2° from 16.5 GHz to 54.7 GHz. The IRR is higher than 40 dB from 21.5 GHz to 53.5 GHz. The input return loss is better than −10 dB at 22–45 GHz, with signal attenuation varying from 5.74 dB to 7.4 dB (including 1:2 power splitting and loss from passive balun). The results of our proposed I/Q signal generation network validate its ability to provide an ultra-precise, fully-differential quadrature signal with compact chip area, low loss and ultra-wide bandwidth.