# A Chopper Stabilization Audio Instrumentation Amplifier for IoT Applications

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## Abstract

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^{2}. The circuit of the M&NEMS microphone and the amplifier was fabricated and measured. From measurement results over a signal bandwidth of 20 kHz, it achieves a signal-to-noise ratio (SNR) of 77 dB.

## 1. Introduction

## 2. Nano- and Micro-Electro-Mechanical System (M&NEMS) Microphone Design

_{dd}/2 with a servo-loop build around A1 that equate source and sink current through the bridge. The differential voltage V

_{in}flowing through the instrumentation amplifier is proportional to the gauge resistance imbalance induced by acoustic vibrations. The power consumption of the electronic circuit is fixed by the current biasing. In the simplest case, the same power supply can bias both sensor and amplifier. In this case, the equivalent resistance R

_{E}of the four gages is quite high to maintain a low-power consumption. Moreover, the supply voltage of the complete M&NEMS nanowire sensor is sufficiently small. However, all current integrated circuits in the industry generally operate with supply voltages greater than 1.2-V. This minimum supply voltage of the M&NEMS nanowire sensor generates a bias I

_{bias}of about 279-μA. This bias current corresponds to a power supply of about 335-μW, which is excessive, on the one hand, because the nanowires cannot dissipate it, and on the other hand, it leads to an increase of the total power consumption that is not compatible with the target IoT application. Therefore, it is necessary to add a circuit that controls the voltage bias of the sensor independently of the amplifier supply voltage. The sensor output common-mode voltage might cause problems for the amplifier when the circuit is referenced at 0 V. In fact, for a 100 μA I

_{bias}current, the bias voltage V

_{bias}is 0.4 V. The sensor common-mode voltage V

_{CM}is about 0.2 V in this case. This V

_{CM}value is assuming the same gauges. Therefore, the amplifier input signals V

_{A}and V

_{B}evolve around an average value of 0.2 V. With the objective of driving this V

_{CM}voltage to a value compatible with that of the amplifier without generating an excessive bias current, it is necessary to control the current of the V

_{CM}voltage independently, and therefore to control V

_{bias}as shown in Figure 4.

_{bias}is one of the key parameters of sensitivity as expressed in Equation (1). It can also be considered that at constant sound pressure, the sensitivity affects the power of the output signal. If the V

_{bias}increases, the output voltage also increases. Therefore, the polarization affects the SNR. However, the current flow through the sensor implies power consumption, which is a crucial parameter in battery-powered systems. Therefore, to optimize the sensor polarization, it is relevant to evaluate its SNR according to its power consumption. The impact of the bias voltage on the SNR is evaluated with the noise model. The thermomechanical noise occurs upstream of the bridge. From the models, the power P

_{signal}of the useful signal is related to the voltage V

_{bias}as:

^{2}

_{Total}denotes the total output noise power of the sensor. Therefore, the power consumption P

_{absorb}depends on the nominal value R

_{0}of one nanogauge and can be written as:

## 3. Low-Noise Instrumentation Amplifier Implementation

_{N}. The voltage V

_{error}is the error voltage that can be written as:

_{error}= 0 V, then:

_{bias}across the nanogauges bridge is controlled via the voltage V

_{bias}such that:

_{N}to cancel the common residual mode voltage V

_{CM}generated at the input of the amplifier when there is an imbalance between I

_{P}and I

_{N}. As a result, at equilibrium, the currents I

_{P}and I

_{N}are identical. The accuracy required depends on the common-mode tolerance of the amplifier.

_{bias}to master the current that crosses the sensor, and thus its sensitivity. I

_{bias}is a lever for acting on sensitivity.

_{A}–V

_{B}. It is composed of amplifiers A1, A2 and A3 as shown in Figure 7. The architecture of each amplifier is based on a two-stage operational transconductance amplifier (OTA) as shown in Figure 8.

_{A}–V

_{B}. It is composed of amplifiers A1 and A2 whose intrinsic gain is very high. The associated passive elements R

_{F}and R

_{G}determine the voltage gain G

_{V1}of the entire stage such that:

_{s}, referenced to the common-mode voltage. Its gain G

_{V2}depends on the elements R

_{2}and R

_{1}such that:

_{V}of the instrumentation amplifier can be written as:

_{1}= 5.1 kΩ, R

_{2}= 160 kΩ, R

_{F}= 2 kΩ and R

_{G}= 390 Ω. Therefore, the ideal total differential gain G

_{V}is 51 dB. Finally, the sensitivity of the sensor is increased by a factor k comprising the product of the gains of the stages of the amplification chain. This simple structure involves the bias voltages V

_{P}and V

_{N}across the sensor. They define both the current I

_{bias}flowing through it and the common-mode voltage V

_{CM}according to Equations (10) and (11) as:

_{0eq}is the equivalent resistance of the sensor and R

_{n}is the resistance of element n. The common-mode voltage V

_{CM}can be critical for the proper functioning of the sensor and its electronics. It must be located between the supply voltage of the amplifier and the reference voltage to allow the excursion of the output voltages of the sensor without saturating the amplifiers that make up the chain. The value V

_{DD}/2 allows the maximum excursion. Therefore, this choice implies conditions on the voltages V

_{P}and V

_{N}. In the case of a sensor preceded by this amplification, structure supplied with a voltage of 2.5 V, for a polarization current I

_{bias}of 100 μA, with nominal nanogauge resistance R

_{0}of 4300-Ω, the voltage V

_{P}must be 1.365 V, V

_{CM}of 1.25 V and V

_{N}of 1.135 V.

_{os}. Therefore, alternating current (AC) spike is caused by the mismatch between the capacitances due to clock feed-through at the Chopper clocks transition moments. The first modulator M1 rectify this AC current. Therefore, a DC spike current appears at its input. The resulting DC spike current has an average value I

_{offset}of:

_{1}and ΔC

_{2}denote the CHS mismatch parasitic capacitance, V

_{clk}denotes the clock signal magnitude and f

_{CH}denotes the chopping clock frequency, with f

_{CH}= 12-kHz. The chopper series impedance and the input signal source are going through by this noise current. Therefore, it depicts as an input voltage spike. The residual offset V

_{os}resulting from the spike average DC value can be written as:

_{os}depicts the spike average DC value. Moreover, a spike voltage V

_{os}is created in the input of Modulator1. This spike voltage causes a low-frequency interference. To cancel-out this interference, the solution is to create a proper delay Δt between Modulator1 and Modulator2. The proposed instrumentation amplifier CHS technique is shown in Figure 10 and Figure 11. The instrumentation amplifier with its common-mode feedback circuit (CMFB) is located between two modulating clock signals m1(t) and m2(t) with period T. Moreover, we introduce a delay ∆t between the two clock signals m1(t) and m2(t) at the same time. Due to the introduction of the delay ∆t, this technique causes a chopping of the spike signal itself. Therefore, the DC content of the output signal V

_{out}(t) is minimized. The residual output dc offset is completely cancelled if an optimal delay value ∆t

_{opt}exists, which can be written as:

_{in}with R denotes the input resistance and C

_{in}denotes the input capacitance of the amplifier. The major weakness of this technique is the τ itself, which not only depends on the sensor’s source resistance R, but also on the amplifier’s input capacitance C

_{in}.

_{out}(t) then contains, apart from higher order harmonics of the chopping frequency, a DC part or residual offset, which is due to chopping artifacts. To solve this problem, shaping of the spike can be introduced by the addition of a first order low-pass filter with time constant τ

_{c}after the amplifier. We must have T >> τ

_{c}>> τ with T is the period of the square wave signal m1(t). The shape of the time response of the filtered spike is primarily determined by τ

_{c}and independent of the impedance of the connected sensor.

_{opt}has been done in such a way that offset reduction is most effective for a worst-case sensor resistance. For our specific implementation ∆t

_{opt}/τ

_{c}= 0.8 has been chosen. The low-pass filter has a cut-off frequency of 30 kHz. The nominal chopping frequency is 12 kHz.

## 4. Measurement Results

_{Supply}consumed by this part of the circuit as a function of the voltage V

_{biasAI}. The measurement results allow us to know the relationship between the current absorbed by the instrumentation amplifier as a function of the bias voltage V

_{bias}. The curve plotted with experimental data and those obtained with the results of simulations for different models of transistors are shown in Figure 15 for comparison. Simulation results are performed in three processes and temperature corners as TT for Typical-Typical, FF for Fast-Fast and SS for Slow-Slow. The curve observed experimentally is similar to that obtained in the simulation with the SF-transistor model. Therefore, the deviation observed when the current exceeds 160 μA can be attributed to the presence of an underestimated access resistance during the simulation. This first characterization identifies the transistor model closest to that of the transistors integrated into the chip.

_{total}is the total current consumption, V

_{rms}is the RMS (root-mean-square) input-referred noise, U

_{th}is the thermal voltage and BW is the bandwidth of the instrumentation amplifier. From Equation (15), it is clear that the NEF parameter includes almost every performance shown in Table 1, namely the equivalent-input referred noise, the power consumption, the bandwidth and indirectly the PSRR (power-supply rejection ratio) and the CMRR (common-mode rejection ratio). In [18,19], the instrumentation amplifier has a high CMRR and high PSRR. However, it has also a high equivalent-input referred noise at list of about 18 nV/√Hz. In [20,21,22], the measured instrumentation amplifier has a low CMRR and low PSRR. Moreover, it has a worse equivalent-input referred noise at least 36 nV/√Hz. Therefore, this noise level affects drastically the instrumentation amplifier and degrades its performances. As a result, all compared instrumentation amplifiers have an equivalent-input referred noise greater than 18 nV/√Hz. On the other hand, our measured instrumentation amplifier has a high CMRR and high PSRR. Moreover, it has the lowest equivalent-input referred noise of only 12 nV/√vHz. As a result, for the same performances, our instrumentation amplifier has a good tradeoff between the supply voltage, the PSRR and the CMRR. Our circuit achieves a NEF of 3.7, a PSRR of 108 dB and a CMRR of 121 dB. Therefore, it proves a competitive performance compared to the state-of-the-art.

## 5. Conclusions

^{2}. The hybrid circuit composed by the M&NEMS microphone and the instrumentation amplifier was fabricated and measured. From measurement results over a signal bandwidth of 20 kHz, the low-noise instrumentation amplifier achieves an SNR of 77 dB and it has a great potential of being used in IoT applications.

## Author Contributions

## Funding

## Acknowledgments

## Conflicts of Interest

## References

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**Figure 1.**Scanning electron microscope (SEM) image of nano- and micro-electro-mechanical system (M&NEMS) nanowire sensor [10].

**Figure 2.**Microphone cross-section with sensing elements and acoustic configuration [11].

**Figure 3.**Microphone top view with the nanogauge and the micro-beams [12].

**Figure 12.**Chip microphotograph of the instrumentation amplifier in CMOS 65-nm technology and 1 mm

^{2}area.

**Figure 13.**Measurement setup of the hybrid circuit composed by the M&NEMS microphone and the ΔΣ modulator.

**Figure 14.**M&NEMS microphone output versus frequency at sound source level of 94 dB in the 20 Hz–20 kHz frequency range.

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**MDPI and ACS Style**

Nebhen, J.; Ferreira, P.M.; Mansouri, S.
A Chopper Stabilization Audio Instrumentation Amplifier for IoT Applications. *J. Low Power Electron. Appl.* **2020**, *10*, 13.
https://doi.org/10.3390/jlpea10020013

**AMA Style**

Nebhen J, Ferreira PM, Mansouri S.
A Chopper Stabilization Audio Instrumentation Amplifier for IoT Applications. *Journal of Low Power Electronics and Applications*. 2020; 10(2):13.
https://doi.org/10.3390/jlpea10020013

**Chicago/Turabian Style**

Nebhen, Jamel, Pietro M. Ferreira, and Sofiene Mansouri.
2020. "A Chopper Stabilization Audio Instrumentation Amplifier for IoT Applications" *Journal of Low Power Electronics and Applications* 10, no. 2: 13.
https://doi.org/10.3390/jlpea10020013