A High-Efficiency QR Flyback DC–DC Converter with Reduced Switch Voltage Stress Realized by Applying a Self-Driven Active Snubber (SDAS)

Abstract

In this paper, a QR flyback converter using a self-driven active snubber (SDAS) was proposed to solve the problem of voltage surge in the switch of QR flyback converters. In the proposed converter, the SDAS consisting of a clamping capacitor and an active switch can be configured in parallel with the main switch or transformer to reduce the voltage surge in the switch. To confirm the steady-state characteristics of the QR flyback converter to which the proposed SDAS is applied, equivalent circuits for each state were constructed, and the equations and characteristics for each state were determined. A 60 W class small AC–DC adapter was constructed to confirm the effectiveness of the proposed converter and the control circuit method, and the experimental results were analyzed. The size of the experimental AC–DC adapter was $74\times 29\times 23 \mathrm{mm}$, and it had a high power density of $20 \mathrm{W}/{\mathrm{in}}^3$ or more. The experimental circuit was limited to the high power conversion efficiency of up to 91.56%, and the maximum voltage surge in the switch was approximately 450 V. One of the reasons for such high efficiency is the SDAS circuit, which sufficiently reduces the voltage surge of the QR flyback switch, compared with the RCD clamp circuit, and does not consume power in principle.

1. Introduction

Recently, due to the rapid development of the IT field, low-power energy consumption and high-efficiency power conversion devices are required. Following the trend of miniaturization and lightweight electronic devices, research is being actively conducted to increase the efficiency of power converters and reduce their weight. To reduce the size and weight of power supply devices, high-frequency switching has become an important design criterion. A switching power supply is a device that converts an input voltage into a stable output voltage, and appropriate circuit development is required depending on the application field and output capacity. For output powers of less than 150 W, flyback converters are most commonly used to achieve miniaturization and low manufacturing cost. In addition, since a transformer is used, multiple outputs can be comprised, and input and output isolation is possible [1,2].

Recently, with the introduction of the quasi-resonant (QR) flyback control method, the switching loss is reduced, and efficiency is increased compared with conventional flyback converters, resulting in a broader range of applications. The power conversion efficiency can be increased by reducing the turn-on loss to reduce the switching loss. However, in the case of an isolated switching power supply such as a QR flyback converter, a turn-off surge in voltage simultaneously occurs across the switch due to internal parasitic components. This high-voltage surge increases the rated voltage of the switch and becomes a major cause of deterioration in the reliability of the product. To reduce high-voltage switch surges, an RCD-structured snubber using a resistor–capacitor–diode can be used across the primary and secondary sides of the transformer or the switch. The RCD snubber has the advantage of having a simple structure and easy design, but it also generates heat in the resistance element and lowers the power conversion efficiency of the converter because the entire energy resulting from the high surge in voltage is consumed in the resistor.

Another way to reduce the voltage surge in the switch is to apply an active clamp circuit by configuring the auxiliary switch in series with the clamp capacitor separately from the main switch. In this case, one of the methods to clamp the main switch voltage to a constant voltage is by introducing a signal in the drive signal of the main switch that is opposite to that of the auxiliary switch. However, this active clamp method cannot be applied to all flyback converter methods and only applies to converters operating in the boundary current mode (BCM), in which the transformer magnetizing inductor current is at the boundary at zero but not applicable to the quasi-resonant (QR) flyback method, which operates in discontinuous conduction mode (DCM) [3,4].

Figure 1 shows the basic circuit of a QR flyback converter with an RCD snubber. Figure 1a shows a converter in which the secondary side of the transformer is composed of a diode rectifier, while Figure 1b shows a converter composed of a synchronous rectifier circuit using MOSFET instead of a diode. In Figure 1a, the main switch of the converter is $S$, and the resistance $R_s$, capacitor $C_s$, and diode $D_s$ of the RCD snubber are in parallel with the transformer $T_1$, and the surge energy due to the resonance of the transformer leakage inductor $L_K$ and the switch parasitic capacitor $C_R$ is consumed in the RCD snubber circuit. In Figure 1b, the main switch of the converter is $S_1$, and the synchronous rectifier switch $S_R$ replaces the diode rectifier on the secondary side of the transformer $T_1$. Figure 2 shows the operating waveform in a steady state. Figure 2a is the voltage and current waveform of the diode rectifier, and Figure 2a is the driving signal and the voltage and current waveform of the MOSFET synchronous rectifier. In the QR flyback converter, the magnetizing current of the transformer increases linearly from zero during the turn-on period of the switch because it operates at the DCM and the continuous conduction mode (CCM). During the turn-off period, a voltage surge is generated during damping resonance due to the transformer leakage inductor and the parasitic capacitor of the switch. Although the RCD snubber circuit absorbs some of the surge current, the remaining current still generates a voltage surge in the switch. At this time, the voltage surge of the switch reaches maximum point, and the voltage surge can cause breakdown of the switch [5]. The secondary-side rectifier current linearly decreases, and after it becomes zero, the resonance resulting from magnetizing inductance $L_M$ and the switch parasitic capacitor $C_R$ starts, and the main switch is turned on at the valley, where the switch resonance voltage is lowest owing to the QR flyback control signal. In the case of the synchronous rectifier, the synchronous rectifier switch $S_R$ is turned on while the rectifier current flows. For the synchronous rectifier switch $S_R$, a MOSFET with low on-resistance is selected to reduce the conduction loss. As shown in Figure 2, since the main switch voltage of the QR flyback converter is turned on when it reaches the lowest point due to resonance, the turn-on loss is reduced, thus reducing switching loss and enabling high-efficiency power conversion. However, since a voltage surge is generated when the switch $S$ is turned off, this problem cannot be fundamentally solved by adding an RCD snubber circuit [6,7,8].

In this paper, a QR flyback converter using a self-driven active snubber (SDAS) is proposed to mitigate the problem of the high switch-turn-off voltage in the QR flyback converter. The proposed converter clamps the voltage surge of the switch to a constant voltage during turn-off operation owing to the SDAS composed of a clamping capacitor and an active switch, and since the active switch drive signal is controlled by the secondary-side current of the transformer, a separate control circuit is not required. In addition, the proposed converter has the advantage of using a switch with a low voltage rating due to the reduced turn-off voltage surge and using the conventional QR flyback control IC. For the design of its major components, the steady-state operation characteristics were analyzed, and the simulation and experimental results were compared. To confirm the effectiveness of the proposed converter and the control circuit method, a 60 W class miniaturized AC–DC adapter was constructed, and the experimental results were analyzed. The size of the experimental AC–DC adapter was $74\times 29\times 23 \mathrm{mm}$, the power density was about $20 \mathrm{W}/{\mathrm{in}}^3$, and the target maximum efficiency was over 90% [9,10,11].

2. QR Flyback DC–DC Converter with SDAS

2.1. Basic Circuit Structure of QR Flyback Converter with SDAS

Figure 3 shows the basic circuit of the QR flyback converter with the SDAS proposed in this paper. In Figure 3a, the SDAS is configured in parallel with the main switch, while in Figure 3b, the SDAS is configured in parallel with the transformer. In this figure, the main switch is $S_1$, and the synchronous rectifier switch $S_R$ replaces the existing diode rectifier on the secondary side of the transformer $T_1$. The SDAS circuit consists of a clamp switch $S_2$ and a clamp capacitor $C_L$ in series. Transformer $T_2$ is a current transformer (CT) that detects secondary-side current and generates driving signals for the synchronous rectifier switch $S_R$ and the clamp switch $S_2$. Therefore, since the two switches $S_R$ and $S_2$ are turned on and off by the secondary-side current signal of the transformer, they are driven independently of the external PWM control signal. The clamp capacitor $C_L$ absorbs the surge energy caused by the resonance of the transformer leakage inductor $L_K$ and the switch parasitic capacitor $C_R$. In the conventional active clamp-type flyback converter, the location of the main switch $S_1$, the clamp switch $S_2$, and the clamp capacitor $C_L$ are the same as the circuit proposed in this paper, but the clamp switch $S_2$ serves as a complementary switch to the main switch, so the QR flyback characteristics are not satisfied. Therefore, the conventional active-clamp flyback converter has different steady-state operation characteristics from the QR flyback converter configured with the SDAS in parallel. Since QR flyback converters are relatively widely applied in the design and manufacturing of small-capacity converters, several manufacturers have released control IC models. In this paper, while maintaining the basic characteristics and design environment of the existing QR flyback converter, we proposed a circuit method capable of high-efficiency power conversion by actively removing the voltage surge of the switch to lower the rated voltage of the switch and reducing the conduction loss of the switch [12,13,14,15,16].

Figure 4 is an equivalent circuit to that in Figure 3a with the transformer $T_1$ removed. Figure 4a is the equivalent circuit with the clamp capacitor $C_L$, and Figure 4b is the equivalent circuit converted to the clamping constant voltage $V_{CL}$ instead of the clamp capacitor. Transformer $T_1$ is represented by the leakage inductor $L_K$ and magnetizing inductor $L_M$, and all switches are assumed to be ideal switches without loss. The resonant capacitor $C_R$ placed in parallel with the main switch $S_1$ is the parasitic capacitor of the switch. The output load resistance and the output capacitor are assumed to be constant voltage in a steady state. In addition, in this study, to simplify the analysis and design, the equivalent circuit in Figure 4a was used as a basis for the steady-state analysis and design of major components [16,17].

2.2. Steady-State Analysis

Figure 5 shows the operating waveform for the equivalent circuit in Figure 4. Figure 5a shows the switch voltage of the converter turned on in the second valley, whereas Figure 5b shows the switch voltage turned on in the first valley. As can be seen, when the converter operates in a steady state, its state is divided into six operating states for one cycle. In this case, the turn-off timing of the switch is different, but the waveform for each state is the same. The operation mechanism in each state of the converter operating under a steady state is described below [18,19].

First, in state 1, the main switch $S_1$ is turned on at time $t_0$, the secondary-side current of the transformer becomes zero, and the auxiliary switch $S_2$ and the synchronous rectifier switch $S_R$ are turned off. The equivalent circuit of state 1 is shown in Figure 6. As can be seen in Figure 5 and Figure 6, the input voltage is applied to the magnetizing inductor $L_M$ and the leakage inductor $L_K$, and the current constantly increases from zero, as shown in Equation (1). Assuming that the main switch voltage is equal to that in Equation (2), and the clamp capacitance is large, the clamp voltage is equal to the turn-off steady-state voltage of the main switch shown in Equation (3). The switch voltage of the synchronous rectifier is determined using Equation (4), and the current is determined with Equation (5). When the main switch $S_1$ is turned off at time $t_1$, the magnetizing current $i_M$ becomes $I_{L1}$, and state 1 ends [20,21].

$$i_{sw}\left(t\right)=i_M\left(t\right)=i_R\left(t\right)=\left(\frac{V_{IN}}{L_R+L_M}\right)t$$

(1)

$$v_{DS1}\left(t\right)=0$$

(2)

$$v_{CL}\left(t\right)=V_{C0}=V_{IN}+N{V}_o$$

(3)

$$N{v}_D\left(t\right)=-\left(V_{IN}+N{V}_o\right)$$

(4)

$$\frac{i_{SR}\left(t\right)}{N}=0$$

(5)

When the main switch $S_1$ is turned off at time $t_1$, state 2 starts, and the equivalent circuit is shown in Figure 7. At time $t_1$, the initial value of the magnetizing current $i_M$ is $I_{L1}$, and the magnetizing inductor $L_M$ and the switch equivalent capacitor $C_R$ resonate, as shown in Equation (6). The switch resonant voltage is determined with Equation (7). At this time, the characteristic impedance $Z_r$ and the resonant angular velocity ${\omega }_r$ are calculated using Equation (8). When the resonance voltage of the main switch $S_1$ becomes equal to the clamp capacitor voltage $V_{CL}$, the time becomes $t_2$, the maximum value of the magnetizing current $i_M$ becomes $I_{L2}$, and state 2 ends.

$$i_M\left(t\right)=i_R\left(t\right)=\left(\frac{V_{IN}}{Z_r}\right)\mathit{sin}{\omega }_{r}t+I_{L1}\mathit{cos}{\omega }_{r}t$$

(6)

$$v_{DS1}\left(t\right)=I_{L1}{Z}_r\mathit{sin}{\omega }_{r}t+V_{IN}\left(1-\mathit{cos}{\omega }_{r}t\right)$$

(7)

$$Z_r=\sqrt{\frac{L_M}{C_R}} {\omega }_r=\frac{1}{\sqrt{L_M{C}_R}}$$

(8)

At time $t_2$, when the magnetizing current $L_M$ flows through the internal diode of the clamp switch, state 3 starts, and the equivalent circuit in this state is shown in Figure 8. At time $t_2$, the initial value of magnetizing current $L_M$ is $I_{L2}$, the magnetizing inductor $L_M$ and the clamp capacitor $C_L$ resonate, and the magnetizing current $i_M$ is determined with Equation (9). The clamp capacitor resonant voltage is provided by Equation (10). At this time, the characteristic impedance $Z_R$ and the resonant angular velocity ${\omega }_R$ are determined using Equation (11). When the clamp current $i_{CL}$ becomes larger than the magnetizing current $i_M$, the magnetizing current becomes $I_{L3}$, the time becomes $t_3$, and state 3 ends.

$$i_{CL}\left(t\right)=i_M\left(t\right)=i_R\left(t\right)=\left(\frac{V_{IN}-V_{C2}}{Z_R}\right)\mathit{sin}{\omega }_{R}t+I_{L2}\mathit{cos}{\omega }_{R}t$$

(9)

$$v_{CL}\left(t\right)=I_{L2}{Z}_R\mathit{sin}{\omega }_{R}t+V_{IN}\left(1-\mathit{cos}{\omega }_{R}t\right)+V_{C2}$$

(10)

$$Z_R=\sqrt{\frac{L_M}{C_L}} {\omega }_R=\frac{1}{\sqrt{L_M{C}_L}}$$

(11)

At time $t_3$, as the resonance current of the clamp current $i_{CL}$ becomes larger than the magnetizing current $i_M$, the remaining current flows through the rectifier on the secondary side of the transformer. When the synchronous rectifier switch $S_R$ and the clamp auxiliary switch $S_2$ are turned on with the rectifier current, state 4 starts, and the equivalent circuit in this state is shown in Figure 9. At time $t_3$, the initial value of the resonance current $i_{CL}$ is $I_{L3}$, the leakage inductor $L_K$ and the clamp capacitor $C_L$ resonate, and the clamp capacitor current is as shown in Equation (12). The clamp capacitor resonant voltage is calculated using Equation (13). The secondary-side current starts at zero due to resonance and performs zero-current switching (ZCS); thus, its waveform is a sinusoidal wave reflecting the resonance. At this time, the characteristic impedance $Z_O$ and the resonant angular velocity ${\omega }_O$ are expressed in Equation (15). When the magnetization current $i_M$ and the clamp capacitor current $i_{CL}$ become equal, the secondary rectifier current becomes zero, and the time becomes $t_4$, and state 4 ends [22,23].

$$i_{CL}\left(t\right)=i_R\left(t\right)=\left(\frac{V_{IN}+N{V}_O-V_{C3}}{Z_o}\right)\mathit{sin}{\omega }_{o}t+I_{L3}\mathit{cos}{\omega }_{o}t$$

(12)

$$v_{CL}\left(t\right)=I_{L3}{Z}_o\mathit{sin}{\omega }_{o}t-\left(V_{IN}+N{V}_O-V_{C3}\right)\mathit{cos}{\omega }_{o}t+\left(V_{IN}+N{V}_O\right)$$

(13)

$$\frac{i_{SR}\left(t\right)}{N}=I_{L3}\left(1-\mathit{cos}{\omega }_{o}t\right)-\frac{N{V}_o}{L_M}t-\left(\frac{V_{IN}+N{V}_O-V_{C3}}{Z_o}\right)\mathit{sin}{\omega }_{o}t$$

(14)

$$Z_o=\sqrt{\frac{L_R}{C_L}} {\omega }_o=\frac{1}{\sqrt{L_R{C}_L}}$$

(15)

At time $t_4$, the magnetizing current $i_M$ and the clamp current $i_{CL}$ become equal, the secondary rectifier current becomes zero, and state 5 begins. The equivalent circuit in this state is shown in Figure 10. When the synchronous rectifier switch $S_R$ and the clamp auxiliary switch $S_2$ are turned off with the rectifier current, the initial value of the resonance current $i_{CL}$ is $I_{L4}$ at time $t_4$. The magnetizing inductor $L_M$ and the clamp capacitor $C_L$ resonate, as shown in the figure, and the clamp capacitor current is calculated with Equation (16). The clamp capacitor resonant voltage is calculated using Equation (17). At this time, the characteristic impedance $Z_R$ and the resonant angular velocity ${\omega }_R$ are expressed in Equation (18). When the magnetizing current $i_M$ and the clamp current $i_{CL}$ become zero, the time becomes $t_5$, and state 5 ends.

$$i_{CL}\left(t\right)=i_M\left(t\right)=i_R\left(t\right)=\left(\frac{V_{IN}-V_{C4}}{Z_R}\right)\mathit{sin}{\omega }_{R}t+I_{L4}\mathit{cos}{\omega }_{R}t$$

(16)

$$v_{CL}\left(t\right)=I_{L4}{Z}_R\mathit{sin}{\omega }_{R}t+V_{IN}\left(1-\mathit{cos}{\omega }_{R}t\right)+V_{C4}$$

(17)

$$Z_R=\sqrt{\frac{L_M}{C_L}} {\omega }_R=\frac{1}{\sqrt{L_M{C}_L}}$$

(18)

At time $t_5$, when the magnetizing current $i_M$, the clamp current $i_{CL}$, and the secondary rectifier current become zero, state 6 starts, and its equivalent circuit is shown in Figure 11. At time $t_5$, the initial value of the turn-off voltage of the main switch is $V_{C5}$, and the magnetizing inductor $L_M$ and the switch equivalent capacitor $C_R$ resonate, as shown in the figure. The magnetizing current is determined using Equation (19), and the switch resonant voltage is expressed in Equation (20). At this time, the characteristic impedance $Z_r$ and the resonant angular velocity ${\omega }_r$ are found using Equation (21). When the switch voltage reaches the lowest point $V_{C6}$ due to resonance, it becomes time $t_6$, and at this time, if the switch is turned on by the control circuit, the turn-on loss can be minimized.

Figure 12 shows a simplified resonant equivalent circuit with the components constituting the resonant circuit for each state equivalent circuit. The initial current and voltage values for each resonant equivalent circuit are shown [24,25].

$$i_M\left(t\right)=i_R\left(t\right)=i_{SW}\left(t\right)=\left(\frac{V_{IN}-V_{C5}}{Z_r}\right)\mathit{sin}{\omega }_{r}t$$

(19)

$$v_{DS1}\left(t\right)=\left(V_{C5}-V_{IN}\right)\mathit{cos}{\omega }_{r}t+V_{IN}$$

(20)

$$Z_r=\sqrt{\frac{L_M}{C_R}} {\omega }_r=\frac{1}{\sqrt{L_M{C}_R}}$$

(21)

To find the appropriate value of the clamp capacitor, the voltage of the magnetizing inductor $L_M$ is obtained using Equation (22). During states 4 and 5, the voltage value is $N{V}_o$, and the maximum current value is $I_M^{+}$. Assuming that the time of state 2–5 is twice the resonance time of the leakage inductor $L_K$ and the clamp capacitor $C_L$, Equation (23) is used. The clamp capacitor value can be obtained from Equation (24). At this time, the maximum voltage value of the clamp capacitor is determined using Equation (25).

$$v_M\left(t\right)=L_M\frac{d{i}_M\left(t\right)}{dt}$$

(22)

$$N{V}_o=L_M\left(\frac{I_M^{+}}{2\times 2\pi \sqrt{L_K{C}_L}}\right)$$

(23)

$$C_L=\frac{1}{L_K}{\left(\frac{L_M{I}_M^{+}}{4\pi N{V}_o}\right)}^2$$

(24)

$$V_{CL}=V_{IN}+N{V}_o$$

(25)

3. Design of QR Flyback DC–DC Converter with SDAS

3.1. Optimal Design Process for Major Devices

The electrical specifications shown in

Table 1. Design conditions for QR flyback converter with SDAS.

Parameter	Symbol	Value	Unit
Input voltage range	$V_{IN}$	90–264	Vac
Output voltage	$V_o$	12	Vdc
Maximum output power	$P_{o max}$	60	W
Maximum output current	$I_{o max}$	5.0	A
Case size	$W\times H\times L$	$29\times 23\times 74$	mm
Efficiency	$\eta$	90	%

were determined for the design of the SDAS-applied QR flyback converter proposed in this paper. The input voltage was AC 90–264 V, the output voltage was set at 12 V, and the maximum output was set at 60 W. Since the AC–DC power supply of less than 75 W is not subject to the IEC 61000-3-2 limitation of current harmonics, the PFC circuit was not used. The size of the power supply device was 29 mm in width, 23 mm in height, and 74 mm in length, and the volume at this time was less than 50 cc.

The winding ratio of the transformer is calculated with Equation (26), and $V_c$ is the voltage that is converted from the output voltage to the primary-side voltage. $K_c$ is the design margin coefficient. In this design, the secondary winding of the transformer was determined as the minimum natural number, so 5 was selected. Since the QR flyback converter always operates in a discontinuous current mode, the entire energy stored in the magnetizing inductor $L_M$ is delivered to the output $P_o$. Therefore, the magnetizing inductor $L_M$ is expressed using Equation (26). The maximum duty rate $d_{max}$ appears at the lowest input voltage and is shown in Equation (28). The RMS values of the current in the primary and secondary windings of the transformer are determined with Equations (29) and (30). Equation (24) can be used for the clamp capacitor, and the result is shown in Equation (31). For the clamp capacitor value, 68 nF was selected, which is the closest standard value [26,27,28,29,30].

$$N=\frac{N_1}{N_2}=\frac{K_c{V}_c}{V_o}=\frac{1.5\times 50}{12}=6.3$$

(26)

$$L_M=\frac{2 P_o}{I_R^2 f_{sw}}=\frac{2\times 70}{{2.5}^2\times 83k}=260 \mathsf{\mu }\mathrm{H}$$

(27)

$$d_{max}=\frac{I_{pk}{L}_M{f}_{sw}}{V_{IN min}}=\frac{2.5\times 269\mu \times 83k}{90\sqrt{2}}=0.42$$

(28)

$$I_{1 rms}=I_{pk}\sqrt{\frac{d_{max}}{3}}=2.5\sqrt{\frac{0.42}{3}}=0.94 \mathrm{A}$$

(29)

$$I_{2 rms}=N{I}_{pk}\sqrt{\frac{1-d_{max}}{3}}=2.5\times 6.3\sqrt{\frac{1-0.42}{3}}=6.93 \mathrm{A}$$

(30)

$$C_L=\frac{1}{5.3\mu }{\left(\frac{260\mu \times 2.5}{4\pi \times 6.4\times 13}\right)}^2=73 \mathrm{nF}$$

(31)

3.2. Circuit Simulation Results

To verify the design results of the QR flyback converter using the SDAS designed in the previous section, a circuit simulation was performed using PSIM 11.0, and the simulation circuit diagram is shown in Figure 13. In the figure, the previously designed values were used for the main components of the converter, and the main switches and transformers were modeled as ideal components. To stabilize the output voltage, a valley-switching PWM control circuit suitable for the QR flyback converter was designed. Figure 14 is the waveform results of the PSIM simulation when the converter operated in a steady state. In Figure 14a, the input voltage was DC 320 V, and the load resistance was 5 Ω; in Figure 14b, the input voltage was DC 320 V, and the load resistance was 2.4 Ω. These values correspond to approximately 30 W and 60 W of output power [31].

In the figure, shown from the top, the operating waveforms refer to the main switch voltage $v_{DS1}$, the magnetizing inductor current $i_M$, the clamp capacitor current $i_{CL}$, the switch drive voltage $v_{GS1,2}$, the transformer secondary rectifier current $i_{SR}$, and the switch current $i_S$. As can be seen from this figure, when the DC input voltage was 320 V, the switch was turned on at the lowest point of the valley. As the power load increased, the number of valleys decreased, and the switch was turned on in the first valley at full load. The driving voltage of the clamp and the synchronous rectifier switch was controlled by the secondary-side rectifier current of the transformer [32]. As a result, it can be seen that the SDAS circuit consisting of the clamp switch and the clamp capacitor operated properly, and the voltage surge was eliminated during the turn-off period of the main switch. Specifically, the rectifier current on the secondary side of the transformer is in the form of a half-wave rectified sinusoidal wave occurred by the resonance, and it showed ZCS characteristics and reduced the switching loss and voltage surge, lowering the rectifier voltage rating [33,34].

In Figure 15, the PSIM simulation results are compared with the experimental results when a 60 W class QR flyback converter with the SDAS was used under a steady state. Shown from the top, the operating waveforms refer to the switch drive voltage $v_{GS1}$, the main switch voltage $v_{DS1}$, the switch current $i_s$, the clamp capacitor current $i_{CL}$, the transformer secondary rectifier current $i_{SR}$, the clamp switch drive voltage $v_{GS2}$, the clamp capacitor voltage $v_{CL}$, and the transformer primary voltage $v_1$. It can be seen that the PSIM simulation results and the experimental waveforms are in relatively good agreement, and they are the same as the operation of each state described above. Therefore, it was confirmed that both the analysis of the steady-state operation and the design process were valid [35].

3.3. Compact AC–DC Adapter Design

Figure 16 shows the basic structure including the control circuit to construct the AC–DC adapter designed above. The basic circuit in the figure is for a QR flyback converter with the SDAS composed of existing commercialized QR flyback control components and an isolation circuit using a photocoupler. As shown in Figure 16, a synchronous rectifier driver component was used to control the synchronous rectifier switch $S_R$, and a photo driver was used to deliver a driving signal to the clamp switch $S_2$ on the primary side of the transformer. As a result, the operation of the synchronous rectifier and the clamp switch was always the same [36,37].

Table 2. Main devices and electrical ratings used in the experimental AC–DC adapters.

Parameter	Name	Model	Specifications	Shape
Main switch	$S_1$	TK31V60X	$600 \mathrm{V}, 30 \mathrm{A}, 78 \mathrm{m}\mathsf{\Omega }$	DFN8 N-ch
Clamp switch	$S_2$	FQP3P50	$-500 \mathrm{V},-2.7 \mathrm{A}, 4.9 \mathsf{\Omega }$	TO-220 P-ch
SR switch	$S_R$	BUK9Y8R5-80E	$80 \mathrm{V}, 100 \mathrm{A}, 6.3 \mathrm{m}\mathsf{\Omega }$	LFPAK56 N-ch
QR controller	$I{C}_1$	NCP1380	$28 \mathrm{V}, 500 \mathrm{mA}$	SO-8
SR controller	$I{C}_2$	TAE1791	$120 \mathrm{V}, 3 \mathrm{A}$	SO-8
Photo driver	$I{C}_3$	TLP118	$3750 \mathrm{V}, 60 \mathrm{ns}$	SO-6 GaAlAs IRED

shows the main components and their electrical characteristics constituting the experimental AC–DC adapter circuit. For the main switch, a MOSFET with a relatively low voltage rating of 600 V was used, as it could reduce the on-resistance compared with other MOSFETs with a similar current rating. A 500 V class P-channel MOSFET was used for the clamp switch and a surface-mounted device (SMD) MOSFET with 6.3 mΩ on-resistance was used for the synchronous rectifier switch to minimize the conduction loss. A QR flyback control IC, NCP1380, was used to control the main switch, TEA1791 was used as the synchronous rectifier driver, and TLP118 was used as the photo driver.

Table 3. Transformer characteristics of experimental AC–DC adapters.

Parameter	Name	Unit	Value	Spec.
Core size	$T_1$		RM7
Primary winding	$N_1$	Turn	32	$0.1 \mathrm{mm}\times 30 \mathrm{p}$
Secondary winding	$N_2$	Turn	5	$0.3 \mathrm{mm}\times 60 \mathrm{p}$
Auxiliary winding	$N_A$	turn	5	$0.1 \mathrm{mm}\times 30 \mathrm{p}$
Magnetizing inductance	$L_M$	$\mathsf{\mu }\mathrm{H}$	260
Leakage inductance	$L_K$	$\mathsf{\mu }\mathrm{H}$	5.3

shows the characteristics of the transformer used in the experimental AC–DC adapter circuit. The magnetic core size of the transformer was TDK’s RM7. For the primary winding, Litz wires comprising a bundle of 30 copper wires with 0.1 mm thickness were used to reduce high-frequency loss, and Litz wires comprising 60 copper wires were used for the secondary winding to deal with a large RMS current value. The measured magnetizing inductance was 260 μH, and the leakage inductance was 5.3 μH. The clamp capacitor was a 68 nF film capacitor rated at 650 V. Figure 17 shows the experimental circuit diagram. In the circuit diagram, the AC–DC rectifier was omitted for convenience, but a general full-wave rectifier was applied in the experiment [38,39,40].

Figure 18 shows the external appearance of the experimental AC–DC adapter. Figure 18a is the assembly drawing of the outer case, and Figure 18b is the inside and outside size and shape of the constructed experimental circuit. The AC–DC adapter was a 74 mm long rectangular cylinder case with a hollow inside and a rectangular lid attached to both ends. The AC terminal was integrated with the lid on one side. The volume of the adapter was 50 cc, and the power density was approximately $20 \mathrm{W}/{\mathrm{in}}^3$. Figure 19 is the instrument composition and test environment to test the experimental circuit. The power supply used in the experiment was a PCR400LE, the electronic load was a PLZ1004WH from Kikusui (Yokohama, Kanagawa, Japan), the power analyzer was a WT1600 from YOKOGAWA (Tokyo, Japan), and the oscilloscope was an HDO6104 model from LeCroy (Chestnut Ridge, NY, USA). To measure the waveform of the experimental circuit in a steady state, the test was carried out without the case.

4. Experimental Results

To verify the effectiveness of the QR flyback converter with the SDAS proposed in this paper, a 60 W AC–DC adapter was tested. Figure 20 is the steady-state operating waveform of the experimental circuit when the input voltage was 310 V. In Figure 20a, the output current was 0.3 A, the voltage $v_{DS}$ of the main switch is shown on the top, and the switch current $i_S$ is shown on the bottom. The power load was approximately 3.6 W, and the switch voltage showed a small duty cycle under a low load current. At this time, the turn-off voltage of the switch was clamped by the SDAS circuit and limited to 420 V. As the load current increased, the operating waveform changed from Figure 20b to Figure 20f. Figure 20f shows when the load current was 5 A, and the output power was 60 W. The maximum switch current was 2.4 A, and the maximum switch voltage was limited to about 450 V. It can be seen from this figure that depending on the load, the peak value of the switch voltage did not significantly increase and was clamped to a constant level [41].

Figure 21 shows a comparison of the switch voltage and current waveforms of a conventional QR flyback converter and the proposed QR flyback converter with the SDAS. The input voltage was 310 V, the load current was 3.5 A, and the output power was 42 W. Figure 21a shows the switch voltage and current waveforms of the experimental circuit using the existing QR flyback converter and RCD snubber, and Figure 21b is the switch voltage and current waveform of the QR flyback converter with the SDAS. As shown in Figure 21a, the voltage surge of the switch occurred at the turn-off moment and had a peak value of up to 550 V. After that, the voltage surge was limited to about 500 V, there was a damping resonance period, and then the voltage surge stabilized at about 410 V, which was the turn-off steady-state voltage. The switch turned on in the second valley, and the switch current had a peak value of about 2 A. At this time, the switch-turn-off voltage also influenced the current during the damping resonance period, and therefore the current resonated simultaneously. As shown in Figure 21b, the turn-off voltage of the switch was clamped by the SDAS circuit and had a peak value of 430 V, and the peak value of the switch current was about 1.9 A. As can be seen in the figure, compared with the RCD clamp circuit, the SDAS circuit reduced the voltage surge of the QR flyback switch by up to 120 V or more. In addition, compared with RCD snubbers, which consume the voltage surge power in resistors, SDAS circuits do not consume power and, therefore, have higher efficiency characteristics in principle [42].

Figure 22 shows the power conversion efficiency of the experimental circuit at an input voltage of 310 V. The efficiency was over 85% when the output power was more than 5 W, and the efficiency was as high as 90% at the maximum power load of 60 W. Figure 23 shows the internal power dissipation of the adapter. At a power load of 10 W, there was a power loss of approximately 1.5 W, and at a maximum power load of 60 W, an internal loss of 6.7 W was observed. Figure 24 shows the output voltage relative to the power load. As evident in the figure, the output voltage remained stable, and a maximum voltage change of less than 5 mV was observed with the change in power load. Figure 25 shows the power conversion efficiency measured at the output power of 20 W, 40 W, and 60 W relative to the input voltage. The efficiency increased at a higher output and with a decrease in the input voltage. Specifically, the maximum power conversion efficiency was achieved when the input voltage was 140 V, and the output was 60 W, at 91.56%. When the output was 60 W, the power conversion efficiency was over 90% in all rates of input voltage.

These experimental results highlight the following findings: The AC–DC adapter of the proposed QR flyback converter with the SDAS showed a high power density of more than $20 \mathrm{W}/{\mathrm{in}}^3$ and a high power conversion efficiency of up to 91.56%. Therefore, the proposed SDAS method can be used as the basic circuit method for high-efficiency power supply. One of the reasons for such high efficiency is due to the SDAS circuit, which sufficiently reduces the voltage surge of the QR flyback switch compared with the RCD clamp circuit and does not consume power in principle. Specifically, a separate control circuit is not required to drive the auxiliary switch for the clamp, and the circuit structure is simplified because it is self-driven inside the converter [43].

5. Conclusions

In this paper, a QR flyback converter using a self-driven active snubber (SDAS) was proposed to solve the problem of the switch voltage surge in the QR flyback converter. In the proposed converter, the SDAS consisting of a clamping capacitor and an active switch was configured in parallel with the main switch or transformer to reduce the voltage surge in the switch. The clamp switch was synchronized with the drive signal of an internal synchronous rectifier and a photo driver or pulse transformer, so no separate control circuit was required.

To confirm the steady-state characteristics of the proposed QR flyback converter with SDAS, equivalent circuits for each state were constructed, and the equations and characteristics for each state were determined. Using the main components for each state, the resonant circuit was separately arranged, and the components involved in resonance and the initial values of each state were identified. The values of the main elements were designed using the steady-state analysis results, and the optimal values of the clamp capacitor were derived. To compare the steady-state analysis results, a simulation using the PSIM program was performed. The simulation results and the theoretical waveforms were relatively well matched, and they showed a very similar shape to the experimental waveforms.

To confirm the effectiveness of the proposed converter and control circuit method, a 60 W class small AC–DC adapter was constructed, and the experimental results were presented. The size of the experimental AC–DC adapter was $74\times 29\times 23 \mathrm{mm}$ and had a high power density of $20 \mathrm{W}/{\mathrm{in}}^3$ or more. The experimental circuit was limited to the high power conversion efficiency of up to 91.56%, and the maximum voltage surge of the switch was approximately 450 V. Taken together, these results highlight the applicability of the proposed converter as the basic circuit method for achieving high-efficiency power supply. One of the reasons for such high efficiency is due to the SDAS circuit, which sufficiently reduces the voltage surge of the QR flyback switch compared with the RCD clamp circuit and does not consume power in principle.